Methods and systems relating to ultra wideband broadcasting

ABSTRACT

Within many applications impulse radio based ultra-wideband (IR-UWB) transmission offers significant benefits for very short range high data rate communications when compared with existing standards and protocols. In many of these applications the main design goals are very low power consumption and very low complexity design for easy integration and cost reduction. Digitally programmable IR-UWB transmitters using an on-off keying modulation scheme on a 0.13 microns CMOS process operating on 1.2V supply and yielding power consumption as low as 0.9 mW at a 10 Mbps data rate with dynamic power control are enabled. The IR-UWB transmitters support new frequency hopping techniques providing more efficient spectrum usage and dynamic allocation of the spectrum when transmitting in highly congested frequency bands. Biphasic scrambling is also introduced for spectral line reduction. Additionally, an energy detection receiver for IR-UWB is presented to similarly meet these design goals whilst being adaptable to address IR-UWB transmitter specificity.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit as a continuation of U.S. patentapplication Ser. No. 15/873,308 filed Jan. 17, 2018, currently pending,which itself claims the benefit of priority from U.S. Ser. No.15/110,489 filed on Jul. 8, 2016, issued as U.S. Pat. No. 10,009,839 on26 Jun. 2018, which itself claims the benefit of priority as a 371National Phase Application of PCT/CA2015/000007 filed Jan. 7, 2015,which itself claims the benefit of U.S. Provisional Patent Application61/925,290 filed Jan. 9, 2014, the entire contents of all beingincorporated herein by reference.

FIELD OF THE INVENTION

This invention relates to ultra wideband wireless communications andmore particularly communications systems exploiting mixerlesstransmitters and energy based receivers.

BACKGROUND OF THE INVENTION

Ultra Wideband (UWB) technology is a wireless technology for thetransmission of large amounts of digital data as modulated codedimpulses over a very wide frequency spectrum with very low power over ashort distance. Such pulse based transmission being an alternative totransmitting using a sinusoidal wave which is then turned on or off, torepresent the digital states, as employed within today's wirelesscommunication standards and systems such as IEEE 802.11 (Wi-Fi), IEEE802.15 wireless personal area networks (PANs), IEEE 802.16 (WiMAX),Universal Mobile Telecommunications System (UMTS), Global System forMobile Communications (GSM), General Packet Radio Service (GPRS), andthose accessing the Industrial, Scientific and Medical (ISM) bands, andInternational Mobile Telecommunications-2000 (IMT-2000).

UWB was formerly known as “pulse radio”, but the Federal CommunicationsCommission (FCC) and the International Telecommunication UnionRadiocommunication Sector (ITU-R) currently define UWB in terms of atransmission from an antenna for which the emitted signal bandwidthexceeds the lesser of 500 MHz or 20% of the center frequency. Thus,pulse-based systems where each transmitted pulse occupies the full UWBbandwidth or an aggregate of at least 500 MHz of narrow-band carrier;for example, orthogonal frequency-division multiplexing (OFDM) can gainaccess to the UWB spectrum under the rules. Pulse repetition rates maybe either low or very high. Pulse-based UWB radars and imaging systemstend to use low repetition rates (typically in the range of 1 to 100megapulses per second). On the other hand, communications systems favorhigh repetition rates (typically in the range of one to two gigapulsesper second), thus enabling short-range gigabit-per-second communicationssystems. As each pulse in a pulse-based UWB system occupies a largebandwidth, even the entire UWB bandwidth, such systems are relativelyimmune to multipath fading but not intersymbol interference, unlikecarrier modulation based systems which are subject to both deep fadingand intersymbol interference (ISI).

Pulse based wireless communication has come a long way since being firstallowed by the Federal Communication Commission (FCC). Able to offereither high data rates or very energy efficient transmissions over shortranges, multiple techniques have been developed for ultra-wideband (UWB)communication including multi-band orthogonal frequency divisionmultiplexing (MB-OFDM), impulse radio (IR-UWB) and frequency modulation(FM-UWB) each with its specific strengths. The potential for very lowpower communications and precise ranging has seen the inclusion of UWBradios in multiple standards aimed for different applications likelow-rate wireless personal area networks (WPAN) with IEEE 802.15.4a andmore recently wireless body area networks (WBAN) with IEEE 802.15.6.

UWB systems are well-suited to short-distance applications in a varietyof environments, such as depicted in FIG. 1 including peripheral anddevice interconnections, as exemplified by first residential environment110, sensor networks, as exemplified by second residential environment120, control and communications, as exemplified by industrialenvironment 130, medical systems, as exemplified by medical imaging 150,and personal area networks (PAN), as exemplified by PAN 140. Due to lowemission levels permitted by regulatory agencies such UWB systems tendto be short-range indoor applications but it would be evident that avariety of other applications may be considered where such regulatoryrestrictions are relaxed and/or not present addressing military andcivilian requirements for communications between individuals, electronicdevices, control centers, and electronic systems for example.

Due to the short duration of UWB pulses in principle it is easier toengineer high data rates and data rate may be exchanged for range inmany instances by aggregating pulse energy per data bit, with theappropriate integration or coding techniques. In addition UWB supportsreal-time location systems and tracking (using distance measurementsbetween radios and precision time-of-arrival-based localizationapproaches) which in addition to its precision capabilities and lowpower make it well-suited for radio-frequency-sensitive environments,such as many medical environments. An additional feature of UWB is itsshort broadcast time.

When considering many applications, such as wireless sensor networks andportable electronics, UWB transceivers should ideally be functionallyhighly integrated for low footprint, support low cost and high volumemanufacturing, and be energy efficient in order to run on a limitedpower source, e.g. a battery, indoor solar cell, small outdoor solarcell, or those developed upon evolving technologies such as thermalgradients, fluid flow, small fuel cells, piezoelectric energyharvesters, micromachined batteries, and power over optical fiber. UWBhas been considered for a long time a promising technology for theseapplications. By using discrete pulses as modulation, it is possible toimplement efficient duty cycling scheme while the transmitter is notactive, see for example Hamdi et al in “A Low-Power OOK Ultra-WidebandReceiver with Power Cycling” (Proc. IEEE New Circuits and SystemsConference 2011, pp. 430-433), which can be further improved by using anOn-Off Shift Keying (OOK) modulation. Further, some UWB operationfrequencies, between 3.1 GHz and 10.6 GHz for example as approved by FCCfor indoor UWB communication systems, see for example “First Report andOrder in the Matter of Revision of Part 15 of the Commission's RulesRegarding Ultra-Wideband Transmission Systems (FCC, ET-Docket 98-153,FCC 02-48), allow for small antennas which can easily be integrated intoan overall reduced footprint sensor solution.

In order to generate very short impulses which conform to a powerspectrum density (PSD) mask, multiple approaches have been attemptedwithin the prior art, each of which has different strengths anddrawbacks. Most work relates to shaping a short numerical impulse byfiltering, see for example Jazairli et al in “An Ultra-Low-PowerFrequency-Tunable UWB Pulse Generator using 65 nm CMOS Technology,”(IEEE Int. Conf. on Ultra-Wideband, 2010, pp.1-4) and Sim et al in “ACMOS UWB Pulse Generator for 6-10 GHz Applications” (IEEE Microwave andWireless Components Letters, Vol. 19(2), pp. 83-85), or by using anoscillator and a mixer to up-convert the signal, see for example Y.Zheng et al., “A 0.18 μm CMOS 802.15.4a UWB Transceiver forCommunication and Localization” (IEEE Int. Solid-State CircuitsConference, 2008, pp. 118-600). However, short impulse filteringrequires bulky passive components and generates a fixed pulse patternwhilst mixing uses an oscillator in conjunction with a mixer with highpower consumption but does provide spectrum flexibility.

Within low power systems controlling the transmitted PSD is veryimportant to maximize the spectrum utilization by appropriately shapingthe pulses. However, in other applications and operating regimesavoiding certain frequency bands may be a requirement in order to reducenoise and the resulting signal interference either to the UWB signal orother signals. For example, global positioning system (GPS) exploit verylow power signals, generally within the noise, at 1575.42 MHz, 1227.60MHz, 1380.05 MHz, 1379.913 MHz, and 1176.45 MHz for the L1 to L5 bandsrespectively, see for example “On the UWB System Coexistence with GSM900, UMTS/WCDMA, and GPS” (IEEE J. Sel. Area in Comms., Vol. 20(9), pp.1712-1721). Whilst mixing can be used for tuning the center frequency ofa transmitter, usually along standardized channels as in IEEE standards,such systems generally use pulses with relatively small bandwidths toseparate the channels, and apart from skipping certain centerfrequencies cannot adaptively adjust their spectral utilisation. Whilstgood spectral usage and tunability may be achieved with MB-OFDM throughthe combination of multiple smaller bandwidth channels concurrently suchapproaches are better suited to high data rate applications due to theincreases in transmitter complexity and power usage.

Accordingly, it would be advantageous for an UWB transmitter to exploitan on-demand oscillator in order to up-convert the pulse therebyremoving the requirement for a separate mixer. It would be furtherbeneficial for the UWB transmitter to be CMOS logic compatible and forthe pulse generation and oscillator to be both digitally tunable inorder to provide control over the pulse bandwidth and center frequencyand capable of rapid frequency adjustments on the order of the pulserepetition rate (PRR). Such UWB transmitters advantageously, incomparison to MB-OFDM UWB transmitters, providing spectralconfigurability, by sequentially changing the transmitted spectrum usinga frequency and bandwidth hopping scheme. It would be further beneficialfor such an UWB transmitter to offer dynamic duty cycling with fastpower up time and OOK modulation to provide reduced power consumption byexploiting the low duty cycle of an IR-UWB symbol and the fact that onlyhalf the symbols require sending energy.

Other aspects and features of the present invention will become apparentto those ordinarily skilled in the art upon review of the followingdescription of specific embodiments of the invention in conjunction withthe accompanying figures.

SUMMARY OF THE INVENTION

It is an object of the present invention to mitigate limitations withinthe prior art relating to ultra wideband wireless communications andmore particularly communications systems exploiting mixerlesstransmitters and energy based receivers.

In accordance with an embodiment of the invention there is provided atransmitter supporting operation as an impulse radio with dynamicfrequency and bandwidth hopping allowing dynamic setting of emittedpower spectrum density.

In accordance with an embodiment of the invention there is provided areceiver supporting operation as an impulse radio receiver with dynamicconfiguration to receive transmitted signals from an impulse radioultra-wideband transmitter.

In accordance with an embodiment of the invention there is provided awireless link comprising a transmitter supporting operation as animpulse radio with dynamic pulse frequency and bandwidth hoppingallowing dynamic setting of emitted power spectrum density, and areceiver supporting operation as an impulse radio receiver with dynamicconfiguration setting to the transmitter.

In accordance with an embodiment of the invention there is provided adevice comprising:

-   a transmitter supporting operation as an impulse radio with dynamic    pulse frequency and bandwidth hopping allowing dynamic setting of    emitted power spectrum density;-   a receiver supporting operation as an impulse radio receiver with    dynamic configuration setting to the transmitter;-   a first power control circuit selectively powering up and powering    down predetermined portions of the transmitter in dependence upon    the data being transmitted; and-   a second power control circuit selectively powering up and powering    down predetermined portions of the receiver in dependence upon the    data being received.

In accordance with an embodiment of the invention there is provided atransmitter supporting operation as an impulse radio with dynamic pulsefrequency and bandwidth hopping allowing dynamic setting of emittedpower spectrum density without up-conversion of the data beingtransmitter.

Other aspects and features of the present invention will become apparentto those ordinarily skilled in the art upon review of the followingdescription of specific embodiments of the invention in conjunction withthe accompanying figures.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the present invention will now be described, by way ofexample only, with reference to the attached Figures, wherein:

FIG. 1 depicts applications of UWB transmitters, receivers, and systemsaccording to embodiments of the invention;

FIG. 2 depicts a block diagram of a UWB transmitter according to anembodiment of the invention;

FIG. 3 depicts a Gaussian pulse generator for a UWB transmitteraccording to an embodiment of the invention;

FIG. 4 depicts a voltage controlled ring oscillator for a UWBtransmitter according to an embodiment of the invention;

FIG. 5 depicts a digitally programmable delay cell for a UWB transmitteraccording to an embodiment of the invention;

FIGS. 6 and 7 depict a programmable variable gain amplifier and driverwith programmable NMOS array for a UWB transmitter according to anembodiment of the invention;

FIGS. 8A and 8B depict the reduced power consumption of a UWBtransmitter according to an embodiment of the invention;

FIG. 9 depicts a design sequence for a fractal antenna together withcompact antenna designs for a UWB transmitter according to an embodimentof the invention;

FIGS. 10 and 11 depict the pulse frequency and pulse width for a UWBtransmitter according to an embodiment of the invention;

FIGS. 12 depicts simulated and measured RF return loss for a fractalantenna a UWB transmitter according to an embodiment of the invention;

FIGS. 13 depicts digitally controllable pulse shape and output powerspectrum density under control tuning for a UWB transmitter according toan embodiment of the invention;

FIG. 14 depicts pulse measurements for a UWB transmitter according to anembodiment of the invention at three different control settings;

FIG. 15 depicts frequency hopping for a UWB transmitter according to anembodiment of the invention;

FIG. 16 depicts pulse amplitude and centre frequency tuning duringfrequency hopping operation of a UWB transmitter according to anembodiment of the invention;

FIG. 17 depicts pulse amplitude and centre frequency tuning duringfrequency hopping operation of a UWB transmitter according to anembodiment of the invention for adjusting PSD at 2.4 GHz;

FIG. 18 depicts pulse amplitude and centre frequency tuning for afrequency hopping UWB transmitter according to an embodiment of theinvention;

FIGS. 19A and 19B depict experimental results for a transmitteraccording to an embodiment of the invention.

FIG. 20 depicts a block diagram of a UWB receiver according to anembodiment of the invention;

FIGS. 21A and 21B depict a system view of a UWB receiver according to anembodiment of the invention;

FIG. 22 depicts an amplifier chain for a UWB receiver according to anembodiment of the invention;

FIG. 23 depicts a single-ended to differential converter withdifferential buffer for a UWB receiver according to an embodiment of theinvention;

FIG. 24 depicts a squaring circuit for a UWB receiver according to anembodiment of the invention;

FIG. 25 depicts differential to single-ended converter for a UWBreceiver according to an embodiment of the invention;

FIG. 26 depicts an integrator for a UWB receiver according to anembodiment of the invention;

FIG. 27 depicts a control signal generating circuit for a UWB receiveraccording to an embodiment of the invention;

FIGS. 28A and 28B depict the reduced power consumption of a UWB receiveraccording to an embodiment of the invention;

FIGS. 29, 30A and 30B depicted experimental results for a receiveraccording to an embodiment of the invention;

FIG. 31 depicts a CMOS integrated circuit implementation for atransmitter and receiver pair according to an embodiment of theinvention;

FIG. 32 depicts a block diagram of a UWB transmitter according to anembodiment of the invention supporting biphasic phase scrambling;

FIG. 33 depicts theoretical spectral profiles for UWB transmitterswithout and with biphasic phase scrambling;

FIG. 34 depicts the pulse shapes for a UWB transmitter employingbiphasic phase scrambling with and without phase shift according to anembodiment of the invention;

FIGS. 35 and 36 depict the pulse frequency and pulse width for a UWBtransmitter employing biphasic phase scrambling according to anembodiment of the invention with increased frequency range to the designproviding similar data in FIGS. 10 and 11 respectively;

FIG. 37 depicts pulse measurements for a UWB transmitter employingbiphasic phase scrambling according to an embodiment of the invention atthree different control settings supporting shorter pulses and higherfrequency operation than the design providing similar data in FIG. 14;

FIG. 38 depicts digitally controllable pulse shape and output powerspectrum density under control tuning for a UWB transmitter employingbiphasic phase scrambling according to an embodiment of the invention;

FIG. 39 depicts pulse amplitude and centre frequency tuning duringfrequency hopping operation of a UWB transmitter employing biphasicphase scrambling according to an embodiment of the invention;

FIG. 40 depicts pulse amplitude and centre frequency tuning for afrequency hopping UWB transmitter employing biphasic phase scramblingaccording to an embodiment of the invention; and

FIG. 41 depicts spectral output shaping of a UWB transmitter employingbiphasic phase scrambling according to an embodiment of the inventionagainst a UWB power-frequency mask.

DETAILED DESCRIPTION

The present invention is directed to ultra wideband wirelesscommunications and more particularly communications systems exploitingmixerless transmitters and energy based receivers.

The ensuing description provides exemplary embodiment(s) only, and isnot intended to limit the scope, applicability or configuration of thedisclosure. Rather, the ensuing description of the exemplaryembodiment(s) will provide those skilled in the art with an enablingdescription for implementing an exemplary embodiment. It beingunderstood that various changes may be made in the function andarrangement of elements without departing from the spirit and scope asset forth in the appended claims.

0. Impulse Radio Ultra Wideband System

As discussed supra UWB offers many potential advantages such as highdata rate, low-cost implementation, and low transmit power, ranging,multipath immunity, and low interference. The FCC regulations for UWBreserved the unlicensed frequency band between 3.1 GHz and 10.6 GHz forindoor UWB wireless communication system wherein the low regulatedtransmitted power allows such UWB systems to coexist with other licensedand unlicensed narrowband systems. Therefore, the limited resources ofspectrum can be used more efficiently. On the other hand, with its ultrawide bandwidth, an UWB system has a capacity much higher than thecurrent narrowband systems for short range applications. Two possibletechniques for implementing UWB communications are Impulse Radio (IR)UWB and multi-carrier or multi-band (MB) UWB. IR-UWB exploits thetransmission of ultra-short (of the order of nanosecond) pulses,although in some instances in order to increase the processing gain morethan one pulse represents a symbol. In contrast MB-UWB systems useorthogonal frequency division multiplexing (OFDM) techniques to transmitthe information on each of the sub-bands. Whilst OFDM has several goodproperties, including high spectral efficiency, robustness to RF andmulti-path interferences. However, it has several drawbacks such as upand down conversion, requiring mixers and their associated high powerconsumption, and is very sensitive to inaccuracies in frequency, clock,and phase. Similarly, nonlinear amplification destroys the orthogonalityof OFDM. Accordingly, MB-UWB is not suitable for low-power and low costapplications.

In contrast IR-UWB offers several advantages, including unlicensed usageof several gigahertz of spectrum, offers great flexibility of spectrumusage, and adaptive transceiver designs can be used for optimizingsystem performance as a function of the data rate, operation range,available power, demanded quality of service, and user preference.Further, multi-Gb/s data-rate transmission over very short range ispossible and due to the ultra-short pulses within IR-UWB it is veryrobust against multipath interference, and more multipath components canbe resolved at the receiver in some implementations, resulting in higherperformance. Further, the ultra-short pulses support sub-centimeterranging whilst the lack of up and down conversion allows for reducedimplementation costs and lower power transceiver implementations.Beneficially, ultra-short pulses and low power transmissions make IR-UWBcommunications hard to eavesdrop upon.

A IR-UWB transmitter as described below in respect of embodiments of theinvention in Section 1 with reference to FIGS. 2 to 19 respectivelyexploits an on-demand oscillator following a pulse generator in order toup-convert the pulses from the pulse generated whilst avoiding therequirement of a separate mixer. Implementable in standard CMOS logicboth the pulse generator and the on-demand oscillator are digitallytunable in order to provide control over the pulse bandwidth and centerfrequency. Further, by exploiting a digitally controlled ring oscillatorfor the on-demand oscillator the IR-UWB transmitter is designed to allowvery quick frequency adjustments on the order of the pulse repetitionrate (PRR). Beneficially this technique provides the same advantages asMB-OFDM in respect of spectrum configurability, achieved by sequentiallychanging the transmitted spectrum using a frequency hopping scheme,whilst maintaining the benefits of IR-UWB. Further, by providingadvanced duty cycling with fast power up time combined with On-Off ShiftKeying (OOK) modulation the IR-UWB according to embodiments of theinvention allows significant reductions in power consumption byexploiting the low duty cycle of a UWB symbol and the fact that onlyhalf the symbols require sending energy.

In addition to defining the operating frequency range for UWB systemsthe different regulatory bodies all specify and enforce a specific powerspectral density (PSD) mask for UWB communications. The PSD maskemployed in respect of embodiments of the invention described below inSection 3 and in respect of FIGS. 20 to 30B is the FCC mask for whichmask data are summarized in Table 1 below for the 3100 MHz-10600 MHz(3.1 GHz-10.6 GHz) range.

TABLE 1 FCC Masks for Indoor - Outdoor for Different Frequency BandsIndoor EIRP Limit Outdoor EIRP Frequency Range (dBm/MHz) Limit (dBm/MHz)<960 −49.2 −49.2 960-1610 MHz −75.3 −75.3 1610-1990 MHz −53.3 −63.31990-3100 MHz −51.3 −61.3 3100-10600 MHz −41.3 −41.3 >10600 MHz −51.3−61.3

Accordingly, it would be evident that the upper limit of −41.3 dB/MHzacross the 3.1 GHz-10.6 GHz frequency range is the same limit imposed onunintentional radiation for a given frequency in order not to interferewith other radios. Basically, for a given frequency, the UWB radiooperates under the allowed noise level which creates the relationshippresented in Equation (1) between E_(p), the transmitted energy perpulse, the maximum spectral power S, the bandwidth B, the bit rate R_(b)and the number of pulses per bits N_(ppb).

E _(p) ·N _(ppb) ·R _(b) ≤S·B  (1)

The IEEE has published a few standards for a physical layer (PHY) forUWB radio in Personal Area Networks (IEEE 802.15.4a-2007), Body AreaNetworks (IEEE 802.15.4a-2007) and Radio-Frequency Identification (IEEE802.15.4f-2012). These standards use mostly relatively large pulsesresulting in relatively narrow bandwidth which is up-converted to aspecific center frequency in order to fill predetermined channels. Thedata is encoded using pulse-position-modulation (PPM) and bi-phasicshift keying (BPSK) is used to encode redundancy data. Every bitconsists of one or more pulses scrambled in phase depending on thetarget data rate. These standards allow considerable flexibility onchannel availability and data rates. The standard also defines thepreamble, headers for the data packet and ranging protocol.

These IEEE standards are designed with multiple users in mind and usedifferent channels to transmit the data, thereby putting a heavyconstraint on pulse bandwidth and limiting the transmitted energy. Priorart on non-standard transmitter attempts to make better use of theavailable spectrum by using narrow pulses, which therefore have a largerbandwidth thereby increasing the maximum transmitted energy according toEquation (1). Accordingly, these transmitters are non-standard and werealso designed for different data rates, frequencies, pulse width, etc.Additionally, they also used various encoding schemes, most notably PPM,OOK or BPSK.

Within this work the inventors have established an energy receiver whichis able to adapt to a variety of IR-UWB pulses and bit encoding therebysupporting communications from both IR-UWB transmitters compliant toIEEE standards as well as those that are non-standard. However, asenergy detection based receivers have no way to extract the phase of thereceived pulses they cannot detect any modulation involving phasedetection such as BPSK. However, most of the other encodings can betranslated in some way into energy levels and proper timing. Thisrequires in turn the ability to achieve synchronization and to adjustthe integration windows appropriately.

Accordingly, an IR-UWB communications link requires a transmittergenerating the ultra-short pulses and a receiver to receive them. Inmany applications these are also co-located as a transceiver. It wouldbe beneficial for these to be implemented using designs compatible withCMOS electronics allowing low cost high volume manufacturing thatleverages existing foundry capabilities as well as allowing the IR-UWBtransceiver to be integrated with additional electronics such as sensorinterfaces, microelectromechanical sensors fabricated through siliconmicromachining and/or post-CMOS processing, microprocessors,microcontrollers, etc.

1. IR-UWB Transmitter Circuit

1A. Transmitter Overview: As depicted in FIG. 2 an IR-UWB transmitter200 according to embodiments of the invention is composed of five mainblocks plus the antenna. First a programmable impulse is produced by apulse generator 230 at clocked intervals when the data signal from ANDgate 210 is high based upon control signals presented to the AND gate210. The pulses from the pulse generator 230 are then up-converted witha programmable multi-loop digitally controlled ring oscillator (DCRO)240, such as that described by Gerosa et al in “A Digitally ProgrammableRing Oscillator in the UWB Range” (IEEE Int. Symp. Circuits and Systems2010, pp. 1101-1104). The output from the DCRO 240 is then coupled to avariable gain amplifier (VGA) 250 in order to compensate for anyfrequency dependency of the pulse amplitude. Finally, a driver 260 feedsthe antenna 270, overcoming typical package parasitics, such as arisingfrom packaging the transceiver within a quad-flat no-leads (QFN)package. In order to further reduce the power consumption of the IR-UWBtransmitter (IR-UWB-Tx) 200 according to embodiments of the invention apower cycling controller 220 dynamically switches on or off thesefunctional blocks when the data signal is low.

1B. Pulse Generator: Now referring to FIG. 3 there is depicted a circuitschematic 300 of a pulse generator 220 according to an embodiment of theinvention implementable in CMOS wherein the pulse generator 220generates a pulse approximating a Gaussian shape when triggered by arising edge of an input Vin. The input is obtained from AND gate 210from a pulse clock and the data signal. This allows full control overthe final symbol shape. The pulse repetition rate (PRR) is determined bythe clock signal frequency and the number of pulses generated depends onthe length of the data signal high time whilst the period of the datasignal determines the data rate.

Initially, the node X is charged to Vdd and Q2 362 is in cutoff. WhenVin goes high, Q1 361 is cut-off and Q3 371 is turned on, transferringthe Vdd level to the input of the first inverter 381. The signal thenpropagates through second to fifth inverters 382 to 385 respectively togenerate the output, Vout, thereby creating the rising edge of theimpulse. When Vout is high, Q2 362 is activated and the node Xdischarges which toggles the first inverter 381. This propagates throughthe inverter chain, comprising second to fifth inverters 382 to 385respectively, lowering Vout thereby creating the falling edge of thepulse. When Vin goes low again between pulses, node X is allowed torecharge, re-arming the pulse generator for the following rising edge ofthe input. To ensure sufficient drive between the inverter chain, firstto fifth inverters 381 to 385 respectively, and the following VCRO 240an output buffer 330 is added at the end.

The pulse width generated is based on the delay through the fiveinverter chain, first to fifth inverters 381 to 385 respectively, ofwhich four are programmable, namely second to fifth inverters 382 to 385respectively. First inverter 381 being coupled to power rail Vdd andground via Q4 341 and Q9 351 whilst second to fifth inverters 382 to 385are coupled to Vdd via first to fourth resistor pairs Q5 to Q8 342 to345 respectively and ground via fifth to eighth resistor pairs Q10 toQ13 352 to 355 respectively. One of the transistors of a pair is alwayson, giving a base current for the longest delay. Turning on the secondtransistor of a pair increases available current to the inverter,thereby reducing its delay. A 4-bit word a0, a1, a2, a3 controls thestarving of the inverters, which may for example be sized to obtain alinearly varying delay with the 4-bit word, such that the bits a0, a1,a2, a3 are coupled to the fifth to eighth resistor pairs Q10 to Q13 352to 355 respectively with the other side of each inverter pair coupled toVdd. The inverse of the 4-bit word, ā0, ā1, ā2, ā3 are coupled to thefirst to fourth resistor pairs Q5 to Q8 342 to 345 respectively with theother side of each inverter pair coupled to Vss. Pulse generator 220 asdepicted by circuit schematic 300 may be powered up and down byenabling/disabling the power rails Vdd and Vss, for example, via gatingtransistors, not shown for clarity, controlled via a power controlsignal or signals.

1C. Programmable Digitally Controlled Ring Oscillator (DCRO): Toup-convert the Gaussian pulse efficiently without requiring either amixer or a phase-locked loop a transmitter according to an embodiment ofthe invention exploits a DCRO 240 as depicted in circuit schematic 400in FIG. 4. In order to allow for toggling of the oscillator by the pulsegenerator, first to fourth transistors Q15 441, Q16 442, Q17 451 and Q18452 have been added in parallel with first to third digitallyprogrammable delay cells (DPDC) 410 to 430 respectively that form thering oscillator. These transistors are on when no input pulse ispresent, which reduces the gain of each DPDC such that the oscillator isoff. The gain of each cell is restored when these transistors are turnedoff by the pulse, allowing the circuit to oscillate. Each DPDC beingcoupled to supply voltage Vdd via Q12 461, Q13 462, and Q14 463 withcontrol line Vpc. The output of the third DPDC 430 is coupled topre-amplifier 440 to generate the output and is again coupled to thesupply rail Vdd via Q19 471 with control line Vpc. Accordingly, the DCRO240 depicted in circuit schematic 400 in FIG. 4 may be selectivelypowered up and down through the Vpc control line.

The maximum oscillation frequency of the DVRO 240 as depicted by circuitschematic 400 in FIG. 4 may be limited by the first to fourthtransistors Q15 441, Q16 442, Q17 451 and Q18 452. Within fabricatedCMOS implementation to date the inventors designs exploiting standardcommercial foundry design rules have yielded oscillation frequencyranges from 3.9 GHz to 9.3 GHz. To equalize the output amplitude overthe frequency range, the pre-amplifier 440 typically drives the nextstage, being VGA 250, with a slanted gain response. According to anembodiment of the invention this may be implanted using a pair ofcascaded common sources with an appropriately sized inductor sized, e.g.the inventors have selected values to induce peaking at approximately 8GHz.

1D. Digitally Programmable Delay Cell: As depicted in FIG. 4 first tothird DPDCs 410 to 430 respectively are employed within the circuitschematic 400 for DVRO 240. Each DPDC of the first to third DPDCs 410 to430 respectively may, for example, be implemented as depicted in circuit500 in FIG. 5 exploiting a latched differential inverter formed by inputtransistor pair 530, comprising Q20 and Q21, which is coupled to outputtransistor pair 540, comprising Q30 and Q31, via a programmable pull-upnetwork comprising pull-up blocks 510. The transistors Q20 and Q30 actas an additional pull-up network to anticipate the switching of theinverter, and compensate for the slower PMOS transistors versus theirNMOS counterparts. This is achieved by driving Q20 and Q30 using theoutput of an earlier cell. Then, that output is used to drive Q21 andQ31, allowing for a reduction in the overall latency of the inverterchain and an increased frequency range.

The latch differential inverter formed by input transistor pair 530 isconnected to eight pull-up blocks 510 comprising first to eighttransistors Q22 to Q29 respectively composed of first and second arrays520A (Q22 to Q25) and 520B (Q26 to Q29) of PMOS transistors whose sizesare binary-weighted. A 4-bit word b0, b1, b2, b3 controls the secondarray 520B whilst the inverse 4-bit word b 0, b 1, b 2, b 3 controls thefirst array 520A. The control bits determined whether the PMOStransistor output of a pull-up block 510 is connected to ground or tothe output node adjusting the drive of the pull-up network. With theaddition of more PMOS transistors connected to the output nodes, thelatching effect is strengthened, requiring an increased time for pulldown and thereby increasing the overall delay of each cell.

1E. Programmable Variable Gain Amplifier: Within the IR-UWB-Tx 200 a VGA250 such as depicted in first circuit 600A in FIG. 6 is implemented tocounteract the frequency dependent amplitude of the DCRO 240 and adjustthe final gain depending on the required output power. As depicted incircuit 600 the VGA 250 is composed of a common source amplifier with anactive load. Capacitor Cb1 601 and resistor Rb 602 are used to decouplethe input of the first circuit 600A and bias the gate of Q33 604.Transistor Q32 605 has a dual functionality, acting as an active loadand cutting off the bias current in the VGA. Accordingly, under thecontrol signal Vpc the bias current may be disabled to Q32 605 or wherethe bias is enabled Q33 605 acts as the active load. To improve theuniformity of the gain over all the operating bandwidth, a seriesinductor L1 606 is added at the output to resonate with the decouplingcapacitor Cb2 607 within the second circuit 600B, Driver 260, at higherfrequencies.

The gain of the first circuit 600A is controlled by a digitallyprogrammable NMOS array (DPNA) 603 which is depicted in FIG. 7 ascomprising a 3:8 bit decoder that activates only one transistor withinthe NMOS Array 720 for each sequence of the 3-bit input control wordc0,c1,c2 . The NMOS Array 720 is composed of eight different sizetransistors which act as a tunable degeneration resistance (r_(0DPNA))and thus control the gain of the VGA 250. The overall voltage gain G_(v)of the VGA 250 being approximately given by Equation (2).

$\begin{matrix}{G \approx \frac{r_{0 - 32}{r_{o - 33}}}{\frac{1}{g_{{mQ}\; 33}} + r_{0\; {DPNA}}}} & (2)\end{matrix}$

1F. Driver: To preserve the up-converted pulse integrity, a driver 260is provided between the VGA 250 output and the 50Ω Antenna 270 andaccount for package parasitics. Second circuit 600B representing adriver 260 according to an embodiment of the invention in FIG. 6 whilstthird circuit 600C represents the package parasitics. As depicted arelatively low Q inductor L2 610 causes a wide frequency peaking toenlarge the frequency of operation of the driver 260. Capacitor Cr 611resonates with L2 610 whilst simultaneously acting as a decouplingcapacitor for the driver 260. Capacitor Ca 612 ensures wideband outputmatching, typically a return loss of at least 10 dB being requiredacross the band of operation. The control transistor Q34 coupled to thepower cycling signal, Vpc, allows the supply to the driver to bedisconnected in order to minimize the power consumption betweentransmitted pulses.

As depicted the third circuit 600C represents the parasitics between thedriver 260 and antenna 270 and comprises a pad capacitance Cpad 616 forthe integrated circuit bond pad together with QFN package resistance,inductance and capacitance represented by R_(QFN) 615, L_(QFN) 613, andC_(QFN) 614 respectively.

1G: Power Cycling: According to embodiments of the invention the lowpower consumption of the IR-UWB-Tx 200 represented by FIGS. 2 to 7respectively is further lowered by the use of the Power CyclingController 220 which as depicted in FIG. 2 is coupled to the PulseGenerator 230, DCRO 240, VGA 250, and Driver 260. The Power CyclingController 220 determines whether to power up/power down these elementsbased upon the data signal and the transmit clock. This Power CyclingController 220 includes, for example, different non-overlapping signalgeneration circuits that create the appropriate signals, such as fortransistors Q12 354 and Q13 355 for Pulse Generator 230 in circuitschematic 300, transistors Q14 463 and Q19 471 for DCRO 240 in circuitschematic 400, transistor Q32 605 in VGA 250 in first circuit 600A inFIG. 6, and transistor Q34 610 in Driver 260 in second circuit 600B inFIG. 6. These signals, generally referred and denoted to as Vpc areappropriately timed by the Power Cycling Controller 220 such that anydisruptions due to the power cycling have minimal impact on the outputsignal. In this manner the inventors have been able to demonstrate thatthe power consumption for an IR-UWB-Tx 200 according to embodiments ofthe invention may be reduced by a factor of 25.

Referring to FIGS. 8A and 8B there are depicted power consumption datafor an exemplary CMOS IR-UWB-Tx according to embodiments of theinvention wherein at full power and maximum transmission rate, using 3pulses per symbol, the IR-UWB-Tx consumes 26.33 mW but in sleep mode0.094 mW, i.e. 94 μW. Accordingly, with power cycling at power on levelsof 10%, 40%, and 70% the IR-UWB-Tx consumes 2.99 mW, 11.35 mW, and 17.53mW respectively. Accordingly, to transmit MPEG-1 the IR-UWB-Tx needoperate at only approximately 7% power cycling and consume approximately2.2 mW. The IR-UWB operates from a 1.2 V supply and measured powerconsumption is as low as 0.9 mW at a 10 Mbps data rate, depending on thefrequency and length of pulses. It is also evident that the powerconsumption with full power cycling varies from approximately 0.84 mW at1 Mbps to approximately 24 mW at 33 Mbps as depicted in FIG. 8B.

1H: Antenna: As depicted in FIG. 2 the output of the driver 260 iscoupled to an antenna 270 to convert the electrical signals within theelectrical circuit to electromagnetic waves propagating through the air.The antenna 270 is the largest component in an UWB system. Varioustechniques have been introduced in an attempt to reduce the footprint ofthe patch antenna. Some of the most promising methods use periodicity inorder to achieve that effect, either by using fractal designs, see forexample Ding et al in “Design of a CPW-fed Ultra Wideband Crown CircularFractal Antenna” (IEEE Int. Symp. Antennas and Propagation, 2006, pp.2049-2052) and Kimar et al in “On the Design of CPW-fed Square OctalShaped Fractal UWB Antenna” (Applied Electromagnetics Conference, 2004,pp. 1-3), or by using a composite Electric-Magnetic-Electric (EME) metalstrip, see for example Chang-Yi et al in “ApplyingElectric-Magnetic-Electric (EME) Composite Metal Strips to Reduce theSize of Patch Antennas” (Asia-Pacific Microwave Conference, 2001, vol.3, pp. 1151-1154). The fractal antenna employed within experimentalmeasurements within this specification is based upon a hexagonaltopology that increases the radiation field with a high current densityat each corner. Referring to FIG. 9 first to fourth images 910 to 940depict the iterative design process with original design, first orderiteration, second order iteration and final antenna respectively. Fifthimage 950 depicts the final antenna which is 14 mm by 16.52 mm andemploys FR4 epoxy substrate with a thickness of 1.6 mm and a dielectricconstant ϵ_(r)=4.4. Two metal strips connected to the ground are addedto the final design in order to increase the bandwidth and improve thereturn loss at low frequencies. The final design takes into account thespecifications of the substrate in optimizing the antenna with respectto the radius of polygons and the slot sizes.

Also depicted in FIG. 9 is an alternate antenna designs depicted aselectrical tracing 960 and first ground plane 970 together with itssimulated and measured electrical return loss in first graph 990 overthe frequency range 10 MHz to 12 GHz indicating operation fromapproximately 3.6 GHz to 11.5 GHz. Similarly polygon fractal antenna 950and alternative antenna 980 are depicted.

2. IR-UWB Transmitter Measurements

2A. Prototyping Board and Test Bench: In order to perform tests on theintegrated circuit and validate its operation, it was necessary todesign a PCB to make a link between the microchip (1.82 mm² dimensions)and the test equipment. The chips were packaged in a 64 pin QFN andconnected to the board through a RF socket designed specifically for thevery small package. The RF output of the transmitter is connected to SMAconnectors with microstrip lines ensuring 50Ω operation at the highfrequencies of operation. To simplify the test setup, all digitalcontrol signals are routed directly out of the chip in parallel. Toconnect all those control signals to the external control system whichis automatically operated by an FPGA (Field -programmable gate array), aVHDCI (very-high-density cable interconnect) was incorporated into thePCB. The VHDCI operates adequately up to 300 MHz which is sufficient forthe digital control signals employed in testing.

Potentiometers were added to each DC voltage to adjust the operatingconditions while allowing offsetting of some of the manufacturingprocess variations. In order to help signal and power integrity,decoupling capacitors between 0.1 μF and 0.1 nF were employed on everynon-RF signal and power pin. The final prototyping board measured 13.1cm×11.9 cm and handles all the various input/output (I/O) signal types;namely the RF input and output of the transceiver, and the controlsignals which are used to adjust the frequency of output pulses (b0 tob3) and its width (a0 to a3). Considering the high number of controlsignals (32 signals for both IR-UWB Transmitter and Receiver), allsignals and clocks are generated with an FPGA Spartan 6. This provides adirect control of the chip by the FPGA, through the VHDCI connectionbetween the two boards. A Logic Analyzer Probe was used to measure allcontrol signals, whilst the RF output was observed with an oscilloscope,e.g. Tektronix Series 70000. This powerful tool allows control of theFPGA directly via a serial communication. This test bench provides afast and efficient setup and presents live observation of the impact ofthe control signal (generated by FPGA) on the transmitter output.

2B. Frequency Modulation: The pulse center frequency was measured forall the bit sequences of the VCRO. As shown in FIG. 10 the frequencyvaries from approximately 2.2 GHz to approximately 4 GHz whilst in FIG.13 the measured pulse shape obtained at three different frequencysettings with their output power spectrum density are depicted.

2C. Pulse Width Modulation: The transmitter pulse width was measured forall bit sequences of the Gaussian pulse generator. Referring to FIG. 11the measured pulse width is depicted a varying from 1120 ps to 2520 psin two operation modes whilst FIG. 14 shows the pulse shape obtained forthree different width settings.

2D. Fractal Antenna: This fractal antenna has been manufactured using aconventional photolithographic process and measured with a VectorNetwork Analyzer using SMA connectors to extract return loss andradiation patterns in an anechoic chamber. The antenna was alsosimulated with the HFSS software suite in order to compare with themeasured parameters. FIG. 12 depicts the simulated and measured returnloss for the fractal antenna 950. The agreement between simulated andmeasured results is excellent. The fractal antenna 950 has an excellentreturn losses performances and radiation pattern, with bandwidth of 8.85GHz, from 3.65 GHz to 12.5 GHz, which is better than the simulationresults.

2E. Spectrum Frequency Hoping: Traditionally, the ability to change thecenter frequency in a UWB transmitter has been used inmulti-channel/multi-band communication systems. These channels allowsmultiple devices to cohabit but also can be selected in order tomitigate interference presents in the environment or avoid specificlocal frequencies. Typically, those channels are well defined and have arelatively narrow bandwidth. This severely limits the amount of energythat can be transmitted because we are using only a fraction of theavailable spectrum. The inventors in contrast have demonstrated a newmethod to maximize the bandwidth efficiency by using a pulse frequencyand bandwidth hopping technique which can be applied to maximize thetransmission power while keeping a fine control on the emitted spectrumto avoid unwanted frequency bands. In order to mitigate the effects ofinterfering signals and maximize the transmitted power the IR-UWB systemaccording to embodiments of the invention provides the ability totransmit several pulses at different frequencies and of various pulseslengths (i.e., pulse bandwidth) thereby providing control of thetransmitted spectrum. Accordingly, IR-UWB transmitters according toembodiments of the invention may exploit frequency and bandwidth hoppingfor both maximizing the spectrum coverage and to avoid interference.Embodiments of the invention may also exploit pulse amplitude hoppingusing the VGA in order to add a degree of freedom to the spectrumconfigurability.

Referring to FIG. 15 there is depicted a pulse sequence from an IR-UWBtransmitter depicting four pulses at 4 GHz followed by three pulses at 3GHz. Similarly referring to FIG. 16 the PSD and pulse profiles for amid-frequency pulse at 3 GHz and a high frequency pulse at 4 GHz aredepicted together with the frequency hopping between operation at 3 GHzand 4 GHz.

2F. Uniform Coverage of the Transmitted Power Spectrum: In order touniformly fill the frequency spectrum between 1.5 GHz and 4.5 GHz, theFPGA was configured to generate a pulse sequence composed of two pulsesat 2.4 GHz with a length of 2.3 ns, followed by three pulses at 3.5 GHzwith a length of 2.38 ns and finally four pulses at 4 GHz with a lengthof 2.5 ns. Sending a different number of pulses for each frequencyallows us to adjust the transmitted power more accurately around thegiven frequency. According, referring to FIG. 18 the benefit of thistechnique of managing the power spectrum density (PSD) of the finalsignal in regards to the spectrum of each individual pulses composingthe frequency hopping sequence can be seen. As depicted the PSD andpulses for sequences at each of the three different frequencies of 2.4GHz, 3.5 GHz, and 4 GHz are depicted together with the 2/3/4 pulsesequence and its resulting PSD which fills the spectrum at around −58dBm over the entire band. It would also be evident to one skilled in theart that the FPGA control could also be integrated within the CMOSApplication Specific Integrated Circuit (ASIC) allowing furtherreductions in the prototype footprint.

The advantage of this approach over traditional filtering of the pulseis an emitted spectrum following the limits more closely. By usingpulses with smaller bandwidth to fill the spectrum we have individualcomponents with a more abrupt fall off, allowing them to be placedcloser to the frequency limits and better filling the mask. It wouldalso be evident that such an approach allows for rapid and simplechanges to the sequence to re-adjust the IR-UWB transmitter to adifferent sub-mask or a different mask without re-designing any elementwithin the system.

2G. Notch in the Transmitted Spectrum: An IR-UWB according to anembodiment of the invention which uses a pulse train with frequencyhopping can be customized to avoid a particular frequency. Thisapplication is similar to a cognitive radio by avoiding any transmissionat frequencies whenever the risk of interference is present. If acommunication system transmits at a frequency within the UWB band,measurements are taken to validate the ability of wide band transmissionpreventing the amplification of the signal at that frequency. To coverthe UWB band and to reduce the risk of interference at an interferingfrequency, pulse sequences are generated with the given frequencycharacteristic. As an example, to avoid transmitting at 2.4 GHz thepulses have the following characteristics; five pulses at 2.2 GHz and2.5 ns length, followed by four pulses at 4 GHz and 2.3 ns length. FIG.17 depicts the PSD and discrete pulse sequences at 2.2 GHz and 4 GHztogether with the frequency hopping results. Also depicted are thecombination pulse sequence and its PSD. Instead of obtaining a powerspectrum density at the same amplitude of −40 dBm over the entire bandwidth (1.5 GHz to 4 GHz); the result of the amplitude of the PSD at 2.4GHz is −55 dBm, a reduction of 15 dB.

2H. Power Spectrum: Referring to FIG. 19A and 19B additional test resultdata for an IR-UWB-Tx according to an embodiment of the invention arepresented. First image 1910 in FIG. 19A depicts the PSD and pulse trainfor the IR-UWB-Tx when hopping between 2.2 GHz and 4 GHz for sequentialpulses whereas second image 1920 shows the direct measurements displayedon the Tektronix Series 70000 oscilloscope for the same frequencyhopping sequence as FIG. 18 with 2 pulses at 2.4 GHz, 3 pulses at 3.5GHz, and 4 pulses at 4 GHz. Referring to FIG. 19B there are depictedfirst to third images 1930 to 1960 representing measured pulses at 2.2GHz, 2.4 GHz, and 4 GHz respectively.

3. IR-UWB Receiver

3A. Receiver Overview: The architecture of an IR-UWB receiver 2000according to an embodiment of the invention is depicted in FIG. 20.Accordingly, the signal from an IR-UWB transmitter is received via anantenna 2010 and coupled to a low noise amplifier (LNA) 2020 followed byfirst amplifier 2030 wherein the resulting signal is squared by squaringcircuit 2040 in order to evaluate the amount of energy in the signal.The output of the squaring circuit 2040 is then amplified with secondamplifier 2050, integrated with integration circuit 2060 and evaluatedby a flash ADC 2070 to generate the output signals. Also depicted isPower Cycling Controller 2080 which, in a similar manner to the powercycling controller 220 of IR-UWB transmitter 200 in FIG. 2, dynamicallypowers up and down the LNA 2020, first and second amplifiers 2030 and2050 respectively, squaring circuit 2040, and flash ADC 2070 to furtherreduce power consumption in dependence of the circuit's requirements.Referring to FIG. 21A the full signal view 2100 of the IR-UWB receiver2000 is depicted whilst FIG. 21B depicts the full signal view of thecontrol signal generator.

Based upon potential applications including, for example, embeddedsensors requiring very low power and low complexity design as well asother power and cost limited system the receiver has to be configurabledigitally using very simple control circuitry. Furthermore, theintegration window has to be easily tunable considering the highsensitivity of energy detection receiver to proper integration windowsynchronization. Different modulation, data rates and burst length willalso need to change the shape of the integration duty cycle and all thepower management must properly keep in synchronization.

3B. Signal Amplification: The first step in the signal path depictedwithin IR-UWB receiver 2000 is an amplification stage comprising LNA2020 and first amplifier 2030. LNA 2010 is designed to match to theantenna allowing for package parasitics. Referring to FIG. 22 withcircuit 2200 then the LNA 2010 is implemented using Q36A 2210 and Q36B2215 whilst first amplifier 2030 comprises first to third common sourcestages Q37 2220, Q38 2230, and Q39 2240 respectively. In each instancecontrol transistors Q40 to Q43 2250 to 2280 respectively are controlledwith control signals CTRL<1> to CTRL<4> respectively whilst each of thefirst to third common source stages Q37 2220, Q38 2230, and Q39 2240respectively are controlled via control signals CTRLp<1> to CTRLp<3>respectively. Ensuring the amplifier chain comprising first to thirdcommon source stages Q37 2220, Q38 2230, and Q39 2240 respectivelymaintains a constant gain over the full 3.1 GHz to 10.6 GHz bandwidthrequires careful management of the gain per stage. Inductive peaking andsource degeneration have been employed by the inventors on all stages tofacilitate reaching the high operating frequency on the 0.13 μm CMOStechnology. By choosing the proper inductors value, inductive peakingcan be adjusted to achieve a nearly flat AC gain on all the stagesleading to an overall flat gain response over the whole UWB bandwidth.

The first stage uses common gate architecture with a cascodedtransistor, see for example Zhang et al in “A Low-Power, Linearized,Ultra-Wideband LNA Design Technique” (IEEE J. Solid-State Circuits, Vol.44(2), pp. 320-330), wherein the load is composed of the inductor forthe high frequency and a resistor to help the lower frequency gain. Alarge PMOS in triode mode is used to provide the required branch currentunder normal operation and cut it during idle time without interferingwith frequency performance and adding only a low serial resistance. Theparasitic capacitance added by the PMOS is included in the value choiceof the load inductor. Another inductor is placed between the cascode andthe common gate NMOS to create a pi network with the parasiticscapacitance allowing these to be neutralized. The inductors used for theentire design described according to an embodiment of the invention aredual layer octagonal coil inductors connected serially. These offer avery small footprint for a given inductor value and a high self-resonantfrequency value at the cost of a lower quality factor (Q). The lower Qis actually desirable as a side-effect by providing a larger bandwidthand a smaller peaking which both contribute to flatten the gain. Thecommon gate architecture low input impedance facilitates matching to the50Ω antenna and has been made to take into account the pads andbondwires parasitics.

The three following amplifier stage are simple common source stages withinductive source degeneration. The loading is provided by a PMOS intriode acting as a resistor for the low frequency gain and an inductorto use peaking to extend the bandwidth and maximal operating frequency.The PMOS double-up as a switch to cut the DC current during the powercycling of the circuit.

3C: Energy Detection: Energy detection with IR-UWB receiver 2000 isachieved through squaring the signal with squaring circuit 2060. Amongstthe simplest methods of squaring a signal is multiplying it by itselfusing a balanced mixer which also has the advantage of a higherlinearity since the even harmonics are cancelled. As a balanced mixerrequires a differential signal in order to create the mixed term, thecorresponding differential signal needs to be created from thesingle-ended input using a single-end input/dual-end output (SEI-DEO)sub-circuit such as depicted with circuit 2300 in FIG. 23.

As depicted a first differential pair 2310 with an unbalanced input actas a single-ended to differential converter (S2D) whilst a seconddifferential pair 2320 act as a differential amplifier acting as abuffer to help scale the currents to drive the squaring circuits largerinput gate capacitances. The S2D uses a capacitor 2330 to generatefeedthrough between the negative output and the fixed common mode inputtransistor to create a pseudo-differential input and improve the phaseand amplitude of the generated differential signals. Inductive loadingof the differential pair is also used to achieve flat gain on the entireUWB spectrum.

Both the unbalanced amplifier and the buffer are designed to be powercycled by modulating the biasing of the tail source. The control signalCTRLp<5> changes the biasing to the ground when the circuit is idle,cutting the biasing current of the differential pair. Since the inputnode take some time to settle after powering up out of the idle state,the feedthrough capacitance has an adverse effect on the settling speedof the circuit. A bypass transistor Q44 2340 driven by CTRLp<4> allowsremoval of the voltage difference during power up, placed in parallel tothe capacitor 2330, helping to settle the outputs to the common DCoutput level.

The mixing circuit 2400 itself is modified from an unloaded doublebalanced Gilbert cell but the output is taken under the current steeringpart. Optionally, any circuit between a squarer or a current redressermay be employed, but squaring helps discriminate the signal from thelower amplitude noise. This configuration represents a compromisebetween a true signal squaring and the output bandwidth. Squaring resultin part of the signal's energy being frequency doubled, which meanproper mixing would require outputting signal component up to 21.2 GHz.Achieving a large bandwidth Gilbert cell mixer running at thesefrequencies would require too much power for most UWB purposes. Usingthe current steering circuit creates a load that is a function of thesignal. Since the difference in the tail current of both currentsteering pair is also a function of the signal where the square value ofthe signal dominates we have an output voltage which is between a fullsquaring and a linear current redresser. The imperfect mixing actuallyhelp keep the operating frequency in a reasonable bandwidth by creatinglot of energy at lower frequency inter-modulation products while stilldiscriminating a stronger signal from the lower power noise. The circuitalso goes into power cycling by cutting the main tail current while idlelike the SEI-DEO sub-circuit depicted as circuit 2300 in FIG. 23.

Finally, a differential-ended input to single-ended output (DEI-SEO)converter as depicted by circuit 2500 in FIG. 25 is used to recover thesingle-ended signal. Within circuit 2500 is a differential pair 2510with an unbalanced output and is followed by a common source amplifier2520 acting as a buffer to regenerate the signal before integration.Common source amplifier 2520 being second amplifier 2050 in IR-UWBreceiver 2000 in FIG. 2.

3D. Energy Integration: As depicted in IR-UWB receiver 2000 followingthe second amplifier 2050 an integrator 2060 is implemented. Referringto FIG. 26 integrating circuit 2600 represents an integrator 2060according to an embodiment of the invention. As depicted integratingcircuit 2600 employs a transconductance amplifier 2610 with parallel RCfeedback to create the integrating circuit 2600. The feedbackcapacitance value can be changed between two settings by the digitalsignal SENS signal acting via Q45 2620 to short the first capacitor C12630 leaving just the second capacitor C1 2640. The two capacitancevalues thus achievable gives two sensitivity settings which can bechosen depending on the strength of the signal. Another transistor Q462650 which is controlled via a SYNC control signals allows the input andoutput to be shorted together, essentially acting as a reset switch.This reset is used to control the integration windows and is held openonly during the actual integration and closed when no pulse burst isexpected. Power cycling is achieved by a PMOS Q47 2660 cutting thecurrent in both branches of the integrator amplifier under action ofCTRL<8>.

Since the integrator is very sensitive to variations on the output andinput during the power up, care has been taken to stabilize the outputvalues and input values during power down by starting the integrator ina known state. Since the signal generated from the energy detectorcircuit is only positive in respect to the analog reference, theintegration direction is strictly positive. By analyzing the outputlevels of the integrator during a training sequence, it is possible totune the reference level to offset the baseline noise power of thecircuit and the background noise in the transmission channel.

3E. Energy Evaluation and Output: The evaluation of the energy level isdone using a 3-bit flash ADC 2070, depicted as being represented by thelast 3 last blocks on the full system view in FIG. 21. The output levelof the integrator can vary from the reference level to V_(DD). Thedesign constraints on the ADC are very low, since the result is morequalitative than quantitative. In order to have feedback on how muchenergy was detected during the integration window the eight discretelevels must be relatively equally spaced and strictly increasing. A goodapproximation of this requirement can be achieved using integratedresistors to generate the proper references voltage for the comparatorsin the flash ADC. The end value is read into flip-flops clocked on theending edge of the integration window signal, allowing the comparator tobe power cycled between readings.

Beside actual detection, having qualitative information on the strengthof the detection value can be used to assess the quality of thedetection and adjust future integration upon the result. The moststraightforward use is to set a detection threshold of energy to accountfor the integrated noise power and reduce the number of false positives.The average noise energy can be easily determined by integrating duringa time where we know there is no data being transmitted, such as forexample during a training sequence. This integrated noise can also beoffset by reducing the integration reference voltage like discussed inthe integrator section.

A minimum detection threshold is very useful, but looking at the maximumintegrated values can also be used to set the integrator sensitivity. Apowerful signal will tend to saturate the integrator quickly, especiallyif multiple pulses are involved per symbol. During longer integrations,reducing the sensitivity of the integrator will increase the absolutedifference between the signal power and the background noise.

In many instances, a training sequence can be employed to achievesynchronization. Unlike transmitters which are aware of the momentswhere there is a transmission or not and can easily power cycle downbetween pulses or symbols, the receiver needs to achieve asynchronization with the transmitter before making full use of the powercycling. Further correct detection of OOK or PPM signal requireslistening exactly during the transmission window of the pulses in orderto assess if pulses where present or not.

Within the prior art research into correct synchronization sequences hasbeen reported and in some instances these sequences are now fixed intothe different UWB standards. When unsynchronized, the receiver canlisten at random intervals and try to detect the synchronizationsequence. Once energy is detected, since the integration is strictlyincreasing, a binary search around the first hit can be done todetermine the timing yielding the most energy. The precision of thesynchronization is determined by the ADC resolution, the length of thetransmitted burst of pulse and is limited by the ability of the basebandcircuit to generate the integration windows. For example, using a 10 nsburst with the 3 bit ADC, the maximum synchronization accuracy is 1.25ns. Achieving the maximum synchronization accuracy depends on theability of the baseband to generate windows with the proper offset.

3F. Configuration, Timing and Energy Management: Within the embodimentsof the invention described within this specification and implemented incurrent prototypes, configuration is performed using a serial datastream. Although reading is done in parallel of all the bits at once inorder to avoid changing configuration while the configuration bits arepushed through the serial line. The data bits include the sensitivity ofthe integrator and various power cycling controls. The IR-UWB receiver(IR-UWB-Rx) circuit has the ability, depending on these bits, toactivate or deactivate power cycling of every component independently oractivate/deactivate it for the whole chip.

IR-UWB-Rx according to an embodiment of the invention exploit anon-overlapping clock generator (NO-ClockGen) 2700 to generate all thepower cycling control signals and the integration window from a singletemplate waveform such that the baseband circuit of the IR-UWB-Rx doesnot need to generate these. This waveform is the only one needed fromthe baseband circuit and is slightly longer than the integration windowitself to accommodate some initial power up time. This ensures that thepower cycling signals and the integration window are always properlyrelated to one another. As depicted NO-ClockGen 2700 comprises firstNAND gate 2710 whose output is coupled to a first array 2740 of delayelements, each of delay D=Δt, to generate the control signalsCTRL<1>,CTRL<2>, CTRL<3>, CTRL<4>, CTRL<5>, CTRL<6>, CTRL<7>after delaysof 2Δt,3Δt,4Δt,5Δt,6Δt,7Δt,8Δt respectively. The output of first NANDgate 2710 after the first delay element in the first array 2740 is fedto an input of second NAND gate 2720 whose output is coupled to a secondarray 2750 of delay elements, each of delay D=Δt, to generate thecontrol signals SYN<1>,SYN<2>, SYN<3>, SYN<4> after delays of3Δt,4Δt,5Δt,6Δt respectively. The output of the second NAND gate 2720 isfed back to the input of the first NAND gate 2710 after a delay of 2Δt .The other inputs of first and second NAND gates 2710 and 2720respectively being coupled to an input via a delay of 2Δt and aninverter 2730 respectively.

All circuits within the IR-UWB-Rx are powered up in sequence to avoidany voltage changes rippling through the circuit and affect theintegrator 2060. To reduce process variation impact, most components aredecoupled and biased independently. However, this can create problemsduring power down where the voltage difference at the decouplingcapacitor is different than during regular operation. The voltagedifference needs to be restored quickly, but most of the circuits havevery high impedance output and inputs. In order to accelerate therecovery from power down, all biasing resistor have a parallel NMOStransistor. These transistors are opened briefly by a pulse CTRLp asindicated in FIGS. 22 to 26 respectively and are generated from thecomponent's CTRL signals edge. When the pulse is active the transistorcreates a lower impedance path to the voltage source. This significantlyhastens the settling time for a little extra logic on the CTRL signal.

4. IR-UWB Receiver Measurements

4.1 Power Consumption: Referring to FIGS. 28A and 28B the powerconsumption of an IR-UWB-Rx according to an embodiment of the inventionis depicted where at full power the IR-UWB-Rx consumes 27 mW whilst insleep mode it consumes approximately 1 μW. Accordingly, with powercycling at power on levels of 10% the IR-UWB-Rx consumes 2.42 mW. Alsoas evident in FIG. 28B the power consumption with full power cyclingvaries from approximately 1 mW at 0.8 Mbps up to approximately 9 mW at7.8 Mbps.

When combining the power consumption data of FIGS. 28A and 28B withFIGS. 8A and 8B for the IR-UWB-Tx then Table 2 summarises the powerconsumption of an IR-UWB transceiver under various conditions.

TABLE 2 IR-UWB Transceiver Power Consumption IR-UWB Mode Tx (mW) Rx (mW)Transceiver (mW) Sleep 0.094 0.001 0.095  10% 2.994 2.428 5.426 100%26.330 26.955 53.285

4.2 Pulse Performance: Referring to FIG. 29 there is depicted themeasured performance of an IR-UWB-Rx according to an embodiment of theinvention whilst FIGS. 30A and 30B depict operation of the IR-UWB undervarying received signal levels. In each instance it can be seen that theoutput is a received bit in response to receipt to a pulse upon therising edge of the Rx_OUT3 signal.

5. IR-UWB Transceiver

Referring to FIG. 31 there are first and second optical micrographs 3110and 3120 for a CMOS IR-UWB transceiver according to an embodiment of theinvention. Third image 3130 depicts the IR-UWB-Tx discretely whilstfourth image 3140 presents the IR-UWB-Tx circuit layout schematic wherethe Pre-Amplifier 440 if the DCRO 240, Driver 260, VCRO (DCRO) 240, VGA250 and Pulse Generator 230—Power Cycling Controller 220 portions areidentified.

Accordingly, it would be evident that embodiments of the invention allowfor low power IR-UWB transmitters with on-demand oscillator allowing anIR-UWB transmitter to exploit spread-spectrum frequency and bandwidthhopping techniques to generate an output PSD conforming to apredetermined regulatory specification and/or mask as well as dynamicmanagement of the PSD to accommodate variations in interference, othertransmitters, etc. Similarly, IR-UWB receivers according to embodimentsof the invention present a low complexity receiver solutionaccommodating IR-UWB transmitters operating with a range of non-phasesensitive protocols.

Further with dynamic power control discrete IR-UWB transmitters, IR-UWBreceivers, and IR-UWB transceivers according to embodiments of theinvention support deployment of personal area networks, body areanetworks, localized electronic device interconnections, etc. within awide range of applications from sensors through to man-machineinterfaces in civil, commercial, and military environments. With lowduty rate powered operation of an IR-UWB receiver a device incorporatingan IR-UWB transceiver may await detection of a wireless “wake” signal.Similarly IR-UWB transmitters and transceivers may dynamically managepower based upon the requirements to transmit data or not as well asfactors such as the required rate and range of the transmitted signals.

6. IR-UWB Transmitter with Biphasic Phase Scrambling

6A. Transmitter Overview: Within the results presented supra in respectof FIGS. 13 and 16-18 the spectral profiles of IR-UWB transmittersaccording to embodiments of the invention contain spectral lines thatare apparent even with frequency scrambling. These spectral lines arepresent within the theoretical simulations performed by the inventors asdepicted in first spectral plot 3300 in FIG. 33. However, the inventorsthrough further theoretical simulations identified that if biphasicphase scrambling is introduced into the IR-UWB transmitter then thespectral lines can be reduced significantly as evident from second image3350 in FIG. 33.

Accordingly, referring to FIG. 32 there is a block diagram 3200 of a UWBtransmitter according to an embodiment of the invention supportingbiphasic phase scrambling. In comparison to the IR-UWB transmitter 200in FIG. 2 for an IR-UWB according to embodiments of the inventionwithout biphasic phase shifting rather than being composed of five mainblocks plus the antenna the Biphasic Phase Shifting IR-UWB (BPS-IR-UWB)transmitter comprises 6 main blocks. First a programmable impulse isproduced by a pulse generator 3230 at clocked intervals when the datasignal from AND gate 3210 is high based upon control signals presentedto the AND gate 3210. The pulses from the pulse generator 3230 are thenup-converted with a programmable multi-loop digitally controlled ringoscillator (DCRO) 3240. The output from the DCRO 3240 is then coupled toa dual-output amplifier (VGA) 3250 both in order to compensate for anyfrequency dependency of the pulse amplitude but also to generate dualphase shifted output signals that are coupled to a switch 3260 whichselects one of the two signals to couple to the output power amplifier(driver) 3280 under the action of the switch control signal “S” appliedto the switch 3260. Note that a similar phase selection scheme could beimplemented by affecting the startup conditions o DCRO 3240 in order toprovide the two phases. This would preclude the need for switch 3260 atthe cost of an added control startup condition control signal on DCRO3240.

The output power amplifier 3280 feeds the antenna 3270, overcomingtypical package parasitics, such as arising from packaging thetransceiver within a quad-flat no-leads (QFN) package. In order toreduce the power consumption of the BPS-IR-UWB transmitter representedby block diagram 3200 according to an embodiment of the invention apower cycling controller 3220 dynamically switches on or off thesefunctional blocks when the data signal “PC” is low. Referring to FIG. 34the pulse shapes for a BPS-IR-UWB transmitter with and without phaseshift are depicted. Accordingly, a BPS-IR-UWB transmitter according toembodiments of the invention transmits pulses with or without phaseshift based upon the control signal “S” applied to switch 3260. If thiscontrol signal is now fed from a random data generator or apseudo-random data generator then the resulting pulses coupled to theantenna of the BPS-IR-UWB transmitter will be pseudo-randomly orrandomly phase shifted.

Referring to FIG. 35 there is depicted the pulse center frequency forall the bit sequences of the DCRO 3240 for a BPS-IR-UWB according to thedesign of the transmitter according to an embodiment of the inventiondepicted in FIG. 32. Due to improvements overall in the prototype UWBtransmitters implemented by the inventors the frequency varies fromapproximately 3 GHz to approximately 7 GHz. Similarly, referring to FIG.36 the transmitter pulse width of a BPS-IR-UWB according to the designof the transmitter according to an embodiment of the invention depictedin FIG. 32 for all bit sequences showing the measured pulse width variesfrom approximately 400 ps to approximately 1400 ps in two operationmodes. FIG. 37 depicts the pulse shape obtained for three differentwidth settings for a UWB transmitter employing biphasic phase scramblingaccording to an embodiment of the invention. The three pulse widthsbeing 0.626 ns, e1.00 ns, and 1.40 ns respectively.

Now referring to FIG. 38 there are depicted the digitally controlledpulse length and associated transmitter output power spectrum densitymeasurements for a PBS-IR-UWB transmitter according to an embodiment ofthe invention under control tuning from a low frequency pulse generation(approximately 3.2 GHz), to a mid-frequency pulse (approximately 4.7GHz), through to a high frequency pulse (approximately 6.0 GHz).Accordingly, as depicted in FIG. 39 the resulting power spectrum andpulse train for a single 3.2 GHz pulse followed by three 6 GHz pulsesunder three different operating conditions of a PBS-IR-UWB according toan embodiment of the invention. Similarly to what is shown in FIG. 17for the IR-UWB embodiment of FIG. 2, FIG. 39 further illustrates theability to omit transmitting within a given frequency band, specifically˜5 GHz in FIG. 39. First and second images 3910 and 3920 respectivelyrepresenting the power spectrum and pulse sequence wherein there is norandom frequency or phase sequencing during the generation andtransmission. Third and fourth images 3930 and 3940 depict the powerspectrum and pulse sequence wherein only a random frequency sequence isemployed such as described supra in respect of an IR-UWB according to anembodiment of the invention such as depicted in FIG. 2 but with thePBS-IR-UWB transmitter according to an embodiment of the invention asdepicted in FIG. 32. In this instance, therefore the switch controlsignal to the switch 3260 within the PBS-IR-UWB is set to one level andmaintained. Finally, fifth and sixth images 3950 and 3960 depict theresults for random frequency and random phase wherein the switch controlsignal to the switch 3260 within the PBS-IR-UWB is fed data to set itslevels from a pseudo-random data generator.

-   -   Accordingly, it would be evident that when comparing first and        third images 3910 and 3930 that the introduction of random        frequency results in reduced spectral lines and that the further        introduction of random phase shifting reduces the spectral lines        even further as depicted in fifth image 3950 and as anticipated        from the theoretical modelling presented and discussed in        respect of FIG. 33. It can also be seen, in fifth image 3950,        that with the pulse sequence of 3.2 GHz and 6 GHz pulses that        the power transmitted between these around 5 GHz is reduced in        comparison to that shown in FIG. 40 image 4050 where pulses at        4.7 GHz are introduced as well.    -   Now referring to FIG. 40 there are depicted the resulting power        spectrum and pulse train for a PBS-IR-UWB according to an        embodiment of the invention frequency hopping within the full        frequency range as depicted in FIG. 35 from approximately 3 GHz        to approximately 7 GHz. This is accomplished using a pulse train        that includes frequencies of 3.2 GHz, (1 pulse) 4.7 GHz (2        pulses) and 6 GHz (3 pulses). First and second images 4010 and        4020 respectively representing the power spectrum and pulse        sequence wherein there is no random frequency or phase        sequencing during the generation and transmission. Third and        fourth images 4030 and 4040 depict the power spectrum and pulse        sequence wherein only a random frequency sequence is employed        such as described supra in respect of an IR-UWB according to an        embodiment of the invention such as depicted in FIG. 2 but with        the PBS-IR-UWB transmitter according to an embodiment of the        invention as depicted in FIG. 32. In this instance, therefore        the switch control signal to the switch 3260 within the        PBS-IR-UWB is set to one level and maintained. Finally, fifth        and sixth images 4050 and 4060 depict the results for random        frequency and random phase wherein the switch control signal to        the switch 3260 within the PBS-IR-UWB is fed data to set its        levels from a pseudo-random data generator. Accordingly, it        would be evident that when comparing first and third images 4010        and 4030 that the introduction of random frequency results in        reduced spectral lines and that the further introduction of        random phase shifting reduces the spectral lines even further as        depicted in fifth image 4050 and as anticipated from the        theoretical modelling presented and discussed in respect of FIG.        33.    -   Now referring to FIG. 41 depicts spectral output shaping of a        UWB transmitter employing biphasic phase scrambling according to        an embodiment of the invention against a UWB power-frequency        mask. In each of first to third images 4110 to 4130 respectively        a pair of UWB masks are depicted establishing a maximum power        level over predetermined frequency ranges as listed in Table 3.

TABLE 3 UWB Masks UWB Mask 1 UWB Mask 2 Frequency Range Max. Signal Max.Signal Band (GHz) Power (dBm) Power (dBm) A 0-950 MHz −49 −49 B 950MHz-1.6 GHz  −75 −75 C 1 1.6 GHz-1.9 GHz −63 2 1.6 GHz-2.0 GHz −53 D 1 1.9 GHz-3.15 GHz −61 2  2.0 GHz-3.15 GHz −51 E 3.15 GHz-10 GHz  −42

Accordingly, first image 4110 represents the power spectrum whereinthere is no random frequency or phase sequencing during the generationand transmission of data. Third image 4120 depicts the power spectrumwherein only a random frequency sequence is employed such as describedsupra in respect of an IR-UWB according to an embodiment of theinvention such as depicted in FIG. 2 but with the PBS-IR-UWB transmitteraccording to an embodiment of the invention as depicted in FIG. 32. Inthis instance, therefore the switch control signal to the switch 3260within the PBS-IR-UWB is set to one level and maintained. Finally, thirdimage 4130 depicts the results for random frequency and random phasewherein the switch control signal to the switch 3260 within thePBS-IR-UWB is fed data to set its levels from a pseudo-random datagenerator. Accordingly, it can be seen that with the reduction ofspectral lines as we progress from first to third images 4110 to 4130respectively that spectrum shaping can be implemented together withspectral line reduction from random phase. As depicted the powerspectrum is compliant with one UWB mask and apart from a couple ofspectral lines below 3 GHz is compliant to the other UWB mask. It isanticipated that adjustment of the frequency hopping and improving thematch between the phase shifted and non-phase shifted signals willfurther reduce the spectral lines and further adjust the power spectrum.

Optionally within other embodiments of the invention the biphasic phaseshifting may be replaced with multiphasic phase shifting (MPS) providingfor a MPS-IR-UWB transmitter although the additional electronic andcontrol complexity may limit application to specific devices and/or UWBapplications.

Specific details are given in the above description to provide athorough understanding of the embodiments. However, it is understoodthat the embodiments may be practiced without these specific details.For example, circuits may be shown in block diagrams in order not toobscure the embodiments in unnecessary detail. In other instances,well-known circuits, processes, algorithms, structures, and techniquesmay be shown without unnecessary detail in order to avoid obscuring theembodiments.

Implementation of the techniques, blocks, steps and means describedabove may be done in various ways. For example, these techniques,blocks, steps and means may be implemented in hardware, software, or acombination thereof. For a hardware implementation, the processing unitsmay be implemented within one or more application specific integratedcircuits (ASICs), digital signal processors (DSPs), digital signalprocessing devices (DSPDs), programmable logic devices (PLDs), fieldprogrammable gate arrays (FPGAs), processors, controllers,micro-controllers, microprocessors, other electronic units designed toperform the functions described above and/or a combination thereof.

The foregoing disclosure of the exemplary embodiments of the presentinvention has been presented for purposes of illustration anddescription. It is not intended to be exhaustive or to limit theinvention to the precise forms disclosed. Many variations andmodifications of the embodiments described herein will be apparent toone of ordinary skill in the art in light of the above disclosure. Thescope of the invention is to be defined only by the claims appendedhereto, and by their equivalents.

Further, in describing representative embodiments of the presentinvention, the specification may have presented the method and/orprocess of the present invention as a particular sequence of steps.However, to the extent that the method or process does not rely on theparticular order of steps set forth herein, the method or process shouldnot be limited to the particular sequence of steps described. As one ofordinary skill in the art would appreciate, other sequences of steps maybe possible. Therefore, the particular order of the steps set forth inthe specification should not be construed as limitations on the claims.In addition, the claims directed to the method and/or process of thepresent invention should not be limited to the performance of theirsteps in the order written, and one skilled in the art can readilyappreciate that the sequences may be varied and still remain within thespirit and scope of the present invention.

What is claimed is:
 1. A receiver circuit supporting operation as animpulse radio receiver having dynamic configuration comprising: a frontend amplification circuit for amplifying received radio frequency (RF)signals from an antenna; a squaring circuit electrically coupled to thefront end amplification circuit for receiving amplified RF signals; anintegration circuit electrically coupled to an output of the squaringcircuit; and an N-bit flash analog-to-digital converter (ADC)electrically coupled to the integration circuit and generating Nmultiple output signals; wherein N is a positive integer and N≥2.
 2. Thereceiver circuit according to claim 1, wherein at least one of: N=3; andan end-value of the N-bit flash ADC is clocked into a series offlip-flops on an ending edge of an integration window signal provided bya control circuit also forming part of the receiver circuit; and theN-bit flash ADC can be power cycled between readings.
 3. The receivercircuit according to claim 1, further comprising a single-ended input todual-ended output sub-circuit disposed between the front endamplification circuit and squaring circuit to generate a pair ofamplified differential signals from the output of the front endamplification circuit which are coupled to the squaring circuit; and adual-ended input to single-ended output sub-circuit disposed betweenoutput of the squaring circuit and the integration circuit to recover asingle-sided output signal from the squaring circuit; wherein the frontend amplification circuit comprises: a low noise amplifier matched tothe antenna for receiving the RF signals from the antenna; a firstamplifier for receiving an output of the low noise amplifier andcomprising a plurality of stages; and the squaring circuit comprises abalanced mixer receiving the differential signals from the single-endedto dual-end sub-circuit comprising a modified unloaded double balancedGilbert cell with a current steering sub-circuit; and the output of thesquaring circuit is established in dependence upon the current steeringsub-circuit.
 4. The receiver circuit according to claim 3, wherein thecurrent steering sub-circuit generates an output voltage as the outputof the squaring circuit which is first function of the squared signalwithin the balanced mixer through the steering sub-circuit creating aload which is a second function of the squared signal and a currentdifference within the current steering sub-circuit which is a thirdfunction of the squared signal; and the output frequency of theelectrical signals from the squaring circuit is less than double that ofthe received RF signals from the front end amplification circuit.
 5. Thereceiver circuit according to claim 1, wherein an integrated referencevoltage of the N-bit flash ADC is adjusted in dependence upon ameasurement of integrated noise power; and the measurement of integratednoise power is established integrating during a time where no data isbeing received.
 6. The receiver circuit according to claim 5, whereinthe time where no data is being received is during receipt of a trainingsequence received by the receiver; and the training sequence allows thereceiver to establish synchronization with a transmitter transmittingthe training sequence.
 7. The receiver circuit according to claim 1,wherein the integration circuit integrates over an integration window;and a sensitivity of the integration circuit is varied in dependenceupon the length of the integration window.
 8. The receiver circuitaccording to claim 1, further comprising a power cycling control circuitfor powering up and down predetermined portions of the receiver in apredetermined order; wherein timing information for the power cyclingcontrol circuit is established in dependence upon synchronization of thereceiver to a transmitter; and synchronization of the receiver with thetransmitter is established in dependence upon the receiver receiving atraining sequence transmitted by the transmitter.
 9. The receivercircuit according to claim 8, wherein the power cycling control circuitemploys a non-overlapping clock generator to generate all power cyclingcontrol signals applied to the front end amplification circuit, squaringcircuit, integration circuit and N-bit flash ADC; and thenon-overlapping clock generator generates the power cycling controlcircuits and an integration window over which the N-bit flash ADCintegrates from a single template waveform.
 10. The receiver circuitaccording to claim 1, further comprising a power cycling control circuitfor powering up and down predetermined portions of the receiver in apredetermined order; wherein a subset of components within the receiverare decoupled and biased independently through a biasing resistor; eachbiasing resistor has a parallel transistor; and each parallel transistoris opened briefly by a control signal pulse provided by the powercycling control circuit in order to create a low impedance path to avoltage source for the biasing resistor associated with the paralleltransistor in order to reduce settling time during powering up of aportion of the receiver circuit comprising a component of the subset ofcomponents associated with the biasing resistor and parallel transistor.11. The receiver circuit according to claim 10, wherein the controlsignal pulse applied to each parallel transistor is generated by a powerup signal associated with the portion of the receiver circuit comprisingthe component of the subset of components to which the paralleltransistor is associated.
 12. A method of receiving data comprising:receiving a plurality of bits of data with an antenna receiving wirelesssignals transmitted by an impulse radio ultra-wideband transmitter, eachbit of data of the plurality of bits of data encoded according to apulse based encoding scheme of a plurality of pulse based encodingschemes comprising a plurality of pulses with frequency and/or bandwidthhopping and a predetermined emitted power spectrum density profile, andprocessing the received plurality of bits with a radio frequency (RF)receiver comprising at least an energy detection circuit; whereinsequential bits of data of the plurality of bits of data employ eitherthe same pulse based encoding scheme of the plurality of pulse basedencoding schemes or a different pulse based encoding scheme of theplurality of pulse based encoding schemes.
 13. The method according toclaim 12, wherein at least one of: the plurality of pulses vary in atleast one of pulse width and pulse amplitude as well as in frequencyand/or bandwidth; and the plurality of pulses comprises a plurality Npulses at a pulse repetition rate established in dependence upon afrequency of clock signal of the impulse radio ultra-widebandtransmitter wherein each pulse of the N pulses is at a predeterminedfrequency of a plurality M frequencies, has a predetermined amplitude,and has a predetermined pulse length; wherein N≥2; M≥2; M and N areintegers; the integer N depends upon a duration of the bit being encodedand transmitted by the impulse radio ultra-wideband transmitter and thepulse repetition rate; and the plurality N pulses are transmitted withinthe duration of the bit of the data signal.
 14. The method according toclaim 12, wherein the radio frequency (RF) receiver further comprises: afront end amplification circuit which amplifies for receiving amplifiedRF signals; and a single-ended input to dual-ended output sub-circuitdisposed between the front end amplification circuit and energydetection circuit to generate a pair of amplified differential signalsfrom the output of the front end amplification circuit which are coupledto the energy detection circuit; and the energy detection circuitcomprises: a squaring circuit electrically coupled to the front endamplification circuit; a dual-ended input to single-ended outputsub-circuit coupled to the squaring circuit to recover a single-sidedoutput signal from the squaring circuit; an integration circuitelectrically coupled to the single-end output of the dual-ended input tosingle-ended output sub-circuit; and an N-bit flash analog-to-digitalconverter (ADC) electrically coupled to the integration circuit andgenerating N multiple output signals; wherein the squaring circuitcomprises a balanced mixer receiving the differential signals from thesingle-ended to dual-end sub-circuit comprising a modified unloadeddouble balanced Gilbert cell with a current steering sub-circuit; andthe output of the squaring circuit is established in dependence upon thecurrent steering sub-circuit.
 15. The method according to claim 12,wherein an integrated reference voltage for the N-bit flash ADC isadjusted in dependence upon a measurement of integrated noise powerwhich is established by integrating during a time where no data is beingreceived.
 16. The method according to claim 12, wherein the RF receiveralso comprises a power cycling control circuit for powering up and downpredetermined portions of the receiver in a predetermined order wheretiming information for the power cycling control circuit is establishedin dependence upon synchronization of the receiver to the impulse radioultra-wideband transmitter through receipt of a training sequence by theRF receiver from the impulse radio ultra-wideband transmitter.
 17. Themethod according to claim 12, wherein the integration circuit integratesover an integration window and a sensitivity of the integration circuitis varied in dependence upon the length of the integration window. 18.The method according to claim 12, wherein the RF receiver also comprisesa power cycling control circuit for powering up and down predeterminedportions of the receiver in a predetermined order where eachpredetermined portion of the RF receiver comprises a component which isdecoupled and biased independently through a biasing resistor inparallel with a transistor to a voltage source, where each transistor isopened briefly by a control signal pulse provided by the power cyclingcontrol circuit in order to create a low impedance path to a voltagesource for the biasing resistor in order to reduce settling time duringpowering up of that portion of the receiver circuit.